IPC-TM-650 EN 2022 试验方法.pdf - 第548页

the substrate extends beyond the circumference of both cav- ity sections. Adjust the separation of the two resonator sec- tions so that the substrate is held by the weight of the upper cavity between the two cavity secti…

5.4.1 Measure the length L of each of the two split-cylinder

resonator sections over several locations and compute the

mean length of both sections.

5.4.2 With the split-cylinder empty (no substrate) and closed

(d=0), find the TE

011

resonance with the network analyzer.

To reduce the coupling losses to a negligible level, adjust

the radial position of the coupling loops so that the peak of

the resonance curve is less than -40 dB. For the particular

10 GHz split-cylinder resonator described in this method, the

resonant frequency should be approximately 10.04 GHz. If

another split-cylinder geometry is being used, use the follow-

ing approximation to estimate the TE

011

resonant frequency of

an empty split-cylinder resonator:

011

2π

√

(

)

(

)

where c is the speed of light in a vacuum, j

is the first zero of

the Bessel function of the first kind J

, a is the split-cylinder

radius in meters and L is the length, in meters, of each of the

split-cylinder sections as shown in Figure 2.

5.4.3 Once the TE

011

resonance has been identified and

displayed on the network analyzer display, measure the reso-

nant frequency f

011

and quality factor Q of the resonance and

use the following expressions to compute the radius a and the

conductivity σ of the empty split-cylinder’s resonator sections:

a=j

[

(

2πƒ

011

)

−

(

)

]

−

σ=

2πƒ

011

where µ

is the permeability of free space and

√

[

(

)

(

)

]

[

(

)

(

)

]

5.5 Estimate the TE

011

Resonant Frequency of

Substrate-Loaded Split-Cylinder Resonator

In addition

to the desired TE

011

resonant mode, other modes are excited

in the split-cylinder resonator as shown in Figure 5. Depend-

ing on the thickness and relative permittivity of the dielectric

substrate being measured, the resonant frequency for the

split-cylinder plus substrate can be significantly lower than the

resonant frequency of the empty split-cylinder resonator as

shown in Figure 6.

In order to identify the correct mode, one can use Figure 6 to

predict the resonant frequency of the TE

011

resonant mode.

For a more accurate estimate of this resonant frequency and

the frequencies of the higher-order resonant modes, software

is available from the National Institute of Standards and Tech-

nology (NIST) which calculates the split-cylinder resonator

dimensions, substrate thickness, and provides an estimate of

the relative permittivity of the substrate. As of the publication

of this method, additional commercial vendors are developing

similar software and will be listed through the IPC-TM-650

Test Methods web page.

5.6 Measure the Relative Permittivity and Loss Tangent

5.6.1

Place the substrate in the gap separating the two cav-

ity sections of the split-cylinder resonator in such a way that

IPC-25513-5

Figure 5 Frequency of the TE

011

Resonant Mode as a

Function of Permittivity and Substrate Thickness for

the 10 GHz Split-Cylinder Resonator

Substrate Thickness (mm)

Sample Relative Permittivity

100

01234

011

Resonant Frequency (GHz)

IPC-TM-650

Number

2.5.5.13

Subject

Relative Permittivity and Loss Tangent Using a Split-Cylinder

Resonator

Date

01/07

Revision

Page3of4

the substrate extends beyond the circumference of both cav-

ity sections. Adjust the separation of the two resonator sec-

tions so that the substrate is held by the weight of the upper

cavity between the two cavity sections.

5.6.2 Using the estimate calculated in 5.3, measure the

resonant frequency and quality factor of the TE

011

resonant

mode using the network analyzer. Since the split-cylinder

resonator is a two-port cavity, the network analyzer should be

set to measure S

, the scattering parameter that measures

the transmission through the cavity. The resonance may have

a significant amount of noise, so it may be necessary to adjust

the amount of averaging performed by the network analyzer.

In some cases where the resonance curve is near the noise

floor, increasing the coupling level of the split-cylinder resona-

tor may be necessary to improve the signal to noise level,

although this may introduce a small amount of coupling loss.

5.6.3 When using the available software, the routine will cal-

culate the relative permittivity and loss tangent of the dielectric

substrate after properly identifying the TE

011

resonant mode.

These values are displayed in the software front panel, includ-

ing an estimate of the measurement uncertainties for the rela-

tive permittivity and loss tangent.

6 Notes If additional measurements are needed at higher

frequencies, the available software will provide the frequencies

of the higher-order TE

0np

resonant modes. The user must

ensure that these modes are symmetric and not distorted by

adjacent resonant modes.

The uncertainties in the real part and loss tangent measure-

ment will be calculated automatically from the uncertainties in

various dimensions that are specified. The major source of

uncertainty will be the uncertainty in the substrate thickness.

Note that the electric field of the TE

011

resonant mode is in the

plane of the substrate. Therefore, if the substrate is anisotro-

pic, the measured component of the relative permittivity also

is in the plane of the substrate.

6.1 Software Availability Software may be available from

commercial vendors and in addition executable code is avail-

able from the Electromagnetic Properties of Materials Project

at the National Institute of Standards and Technology (NIST,

Boulder, CO). As commercial vendor software becomes avail-

able, IPC will provide listings for these at the IPC-TM-650 web

page located at www.ipc.org, under ‘‘Standards.’’

6.2 References

M.D. Janezic, ‘‘Nondestructive Relative Permittivity and Loss

Tangent Measurements using a Split-Cylinder Resonator,’’

Ph.D. Thesis, University of Colorado at Boulder, 2003.

M.D. Janezic, E.F. Kuester, J. Baker-Jarvis, ‘‘Broadband

Complex Permittivity Measurements of Dielectric Substrates

using a Split-Cylinder Resonator,’’ IEEE MTT-S International

Microwave Symposium Digest, pp. 1817-1820, 2004.

M.D. Janezic and J. Baker-Jarvis, ‘‘Full-Wave Analysis of a

Split-Cylinder Resonator for Nondestructive Permittivity Mea-

surements,’’ IEEE Transactions on Microwave Theory and

Techniques, vol. 47, no. 10, pp. 2014-2020, 1999.

K.J. Coakley, J.D. Splett, M.D. Janezic, R.F. Kaiser, ‘‘Estima-

tion of Q-factors and resonant frequencies,’’ IEEE Transac-

tions on Microwave Theory and Techniques, vol. 51, no. 3,

pp. 862-868, 2003.

J. Baker-Jarvis et al, ‘‘Dielectric and Conductor Measure-

ments of Electronic Packaging Materials,’’ NIST Technical

Note 1520, 2001.

J. Baker-Jarvis et al, ‘‘Measuring the permittivity and perme-

ability of lossy materials: solids, liquids, building materials, and

negative-index materials,’’ NIST Technical Note 1536, 2005.

B.N. Taylor, ‘‘Guidelines for Expressing the Uncertainties of

NIST Measurement Results,’’ NIST Technical Note 1297,

1994.

IPC-25513-6

Figure 6 Typical Multiple Split-Cylinder Resonator

Resonances

Frequency (GHz)

-60

-70

-80

-90

8.6 8.7 8.8 8.9 9.0

Magnitude of S

(dB)

IPC-TM-650

Number

2.5.5.13

Subject

Relative Permittivity and Loss Tangent Using a Split-Cylinder

Resonator

Date

01/07

Revision

Page4of4

1 Scope and Purpose

1.1 Scope

This document describes the frequency domain

test methods to accurately determine the amount of signal

propagation loss and delay for electrical printed boards, to

meet the demand of high speed applications nowadays. As

the data rate of high speed IO continues to increase (e.g., 10

Gbps and above), production testing and development testing

require more precise and accurate high frequency methods.

(Existing IPC-TM-650 Test Methods such as Method

2.5.5.12A are not adequate). Additionally, previous IPC test

methods do not encompass traditional industry methods

using VNA, such as thru-reflect-line (TRL), and recent devel-

opments of 2X-Thru test methods, etc. This test method is

defined to close the gaps.

The scope of this test method includes:

• Calibration and/or de-embedding techniques

• Probing/test fixture choices that impact measurement

quality

• Coupon Design

• Test sample pre-conditioning

• Environmental impact, etc.

1.2 Purpose

1.2.1 The importance of Setting up Correct Reference

Plane for Printed Board Characterization

The impor-

tance of setting up a correct reference plane in a typical inter-

connect measurement setup is illustrated in Figure 1-1. The

vector network analyzer (VNA) has been the de-facto standard

for accurate passive interconnect characterization including

the printed circuit board, connector, cables, etc. Making high

quality VNA measurement is straight-forward with standard

coaxial connectors and precision SOLT (short, open, load,

through) calibration kits. However, test fixtures are usually

required to connect the standard coaxial connectors to the

non-coaxial device under test (DUT). SOLT calibration can

readily move the reference plane to Ref plane A and Ref plane

A’ in the figure, while the intended DUT is the printed board

conductor only (between Ref plane B and Ref plane B’). The

test fixtures (between A and B, A’ and B’) need to be charac-

terized and then de-embedded to recover the insertion loss of

DUT.

Microwave probes are often used to probe interconnect struc-

tures for quick measurement, as shown Figure 1-2. A similar

calibration or de-embedding procedure is needed to move the

reference plane to the target location (Ref plane B and B’

shown in the figure). Note that sometimes, an SOLT calibra-

tion procedure can be carried out using calibration substrates

provided by probe vendor, to move the reference plane to the

probe tip, but it does not move the reference plane to the tar-

get location and additional de-embedding procedure is still

needed.

In a general calibration/de-embedding process, specialized

calibration standards with known electrical properties are

inserted at the end of the test fixture, and a calibration pro-

cess is performed to move the reference plane to the end of

IPC-25514-1-1

Figure 1-1 Reference Planes in Printed Board Insertion

Loss Characterization

IPC-25514-1-2

Figure 1-2 Reference Planes in Printed Board Insertion

Loss Characterization with Microwave Probe

3000 Lakeside Drive, Suite 105N

Bannockburn, IL 60015-1249

IPC-TM-650

TEST METHODS MANUAL

Number

2.5.5.14

Subject

Measuring High Frequency Signal Loss and

Propagation on Printed Boards with Frequency

Domain Methods

Date

02/2021

Revision

Originating Task Group

High Frequency Signal Loss Test Methods Task

Group (D-24D)

Material in this Test Methods Manual was voluntarily established by Technical Committees of IPC. This material is advisory only

and its use or adaptation is entirely voluntary. IPC disclaims all liability of any kind as to the use, application, or adaptation of this

material. Users are also wholly responsible for protecting themselves against all claims or liabilities for patent infringement.

Equipment referenced is for the convenience of the user and does not imply endorsement by IPC.

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